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LTC3410 2.25MHz, 300mA Synchronous Step-Down Regulator in SC70
FEATURES

DESCRIPTIO
High Efficiency: Up to 96% Very Low Quiescent Current: Only 26A Low Output Voltage Ripple 300mA Output Current at VIN = 3V 380mA Minimum Peak Switch Current 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle Stable with Ceramic Capacitors 0.8V Reference Allows Low Output Voltages Shutdown Mode Draws < 1A Supply Current 2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Available in Low Profile SC70 Package
The LTC (R)3410 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 26A, dropping to <1A in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3410 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors. The LTC3410 is specifically designed to work well with ceramic output capacitors, achieving very low output voltage ripple and a small PCB footprint. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.8V feedback reference voltage. The LTC3410 is available in a tiny, low profile SC70 package.
, LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131, 5994885.
APPLICATIO S

Cellular Telephones Wireless and DSL Modems Digital Cameras MP3 Players Portable Instruments
TYPICAL APPLICATIO
VIN 2.7V TO 5.5V VIN RUN VFB GND SW
Efficiency and Power Loss vs Output Current
100 1 90
VOUT 2.5V
EFFICIENCY (%)
4.7H CIN 4.7F CER LTC3410 10pF COUT 4.7F CER
80 70 60 50 40 30 20 10 0 0.1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 10 100 OUTPUT CURRENT (mA) POWER LOSS EFFICIENCY
887k 412k
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0.1
POWER LOSS (W)
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0.01
0.001
0.0001 1000
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LTC3410
ABSOLUTE
(Note 1)
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RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN 1 GND 2 SW 3 6 VFB 5 GND 4 VIN
Input Supply Voltage .................................. - 0.3V to 6V RUN, VFB Voltages ..................................... - 0.3V to VIN SW Voltage (DC) ......................... - 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 500mA N-Channel Switch Sink Current (DC) ................. 500mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. - 40C to 85C Junction Temperature (Notes 3, 5) ...................... 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
ORDER PART NUMBER LTC3410ESC6 SC6 PART MARKING LBSD
SC6 PACKAGE 6-LEAD PLASTIC SC70 TJMAX = 125C, JA = 250C/ W
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 3.6V unless otherwise specified.
SYMBOL IVFB IPK VFB VFB VLOADREG VIN VUVLO IS PARAMETER Feedback Current Peak Inductor Current Regulated Feedback Voltage Reference Voltage Line Regulation Output Voltage Load Regulation Input Voltage Range Undervoltage Lockout Threshold Input DC Bias Current Burst Mode(R) Operation Shutdown Oscillator Frequency RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET SW Leakage RUN Threshold RUN Leakage Current VIN Rising VIN Falling (Note 4) VFB = 0.83V, ILOAD = 0A VRUN = 0V VFB = 0.8V VFB = 0V ISW = 100mA ISW = -100mA VRUN = 0V, VSW = 0V or 5V, VIN = 5V

CONDITIONS
MIN 380
TYP 490 0.8 0.04 0.5
MAX 30
UNITS nA mA V %/V % V V V A A MHz kHz A V A
VIN = 3V, VFB = 0.7V, Duty Cycle < 35% VIN = 2.5V to 5.5V ILOAD = 50mA to 250mA

0.784
0.816 0.4 5.5
2.5 2 1.94 26 0.1 1.8 2.25 310 0.75 0.55 0.01 0.3 1 0.01
2.3
35 1 2.7 0.9 0.7 1 1.5 1
fOSC RPFET RNFET ILSW VRUN IRUN
Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3410E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3410: TJ = TA + (PD)(250C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
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LTC3410 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1 Except for the Resistive Divider Resistor Values) Efficiency vs Input Voltage
100 90 80 IOUT = 250mA 70 60 50 40 VOUT = 1.8V 30 4.5 4 2.5 3 3.5 INPUT VOLTAGE (V) IOUT = 0.1mA IOUT = 100mA
100 90 80
EFFFICIENCY (%)
IOUT = 10mA
EFFICIENCY (%)
IOUT = 1mA
60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
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EFFICIENCY (%)
Reference Voltage vs Temperature
0.814 VIN = 3.6V
2.7 2.6
OSCILLATOR FREQUENCY (MHz)
REFERENCE VOLTAGE (V)
2.5 2.4 2.3 2.2 2.1 2.0 1.9
OSCILLATOR FREQUENCY (MHz)
0.809 0.804 0.799 0.794 0.789 0.784 -50 -25
50 25 75 0 TEMPERATURE (C)
OUTPUT VOLTAGE (V)
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5
3410 G02
Efficiency vs Output Current
100 90 80 70 60 50 40 30 20 10
Efficiency vs Output Current
70
5.5
VOUT = 1.8V 0 1 10 100 0.1 OUTPUT CURRENT (mA)
VOUT = 1.2V 0 1 10 100 0.1 OUTPUT CURRENT (mA)
VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000
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Oscillator Frequency vs Temperature
VIN = 3.6V 2.7 2.6 2.5 2.4 2.3 2.2 2.1 2.0 1.9 -25 0 25 50 75 TEMPERATURE (C) 100 125 1.8
Oscillator Frequency vs Supply Voltage
100
125
1.8 -50
2
3 5 4 SUPPLY VOLTAGE (V)
6
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Output Voltage vs Load Current
1.84 1.82 1.80 1.78 1.76 1.74 1.72 1.70 1.68 0 100 200 400 300 LOAD CURRENT (mA) 500
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LTC3410 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1 Except for the Resistive Divider Resistor Values)
RDS(ON) vs Input Voltage
1.2 1.1 1.0 0.9
RDS (ON) ()
1.0 1.2
DYNAMIC SUPPLY CURRENT (A)
0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 1 2
MAIN SWITCH
RDS (ON) ()
SYNCHRONOUS SWITCH
5 4 3 INPUT VOLTAGE (V)
Dynamic Supply Current vs Temperature
50 110 100
DYNAMIC SUPPLY CURRENT (A)
40
SWITCH LEAKAGE (nA)
30
20
10
0 -50
-25
50 25 0 75 TEMPERATURE (C)
4
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RDS(ON) vs Temperature
50
VIN = 4.2V VIN = 2.7V 0.8 0.6 VIN = 4.2V 0.4 VIN = 2.7V 0.2 VIN = 3.6V MAIN SWITCH SYNCHRONOUS SWITCH VIN = 3.6V
Dynamic Supply Current vs VIN
VOUT = 1.2V ILOAD = 0A 40
30
20
10
7
0 -50 -30 -10 10 30 50 70 90 110 130 TEMPERATURE (C)
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0
1
2
3 VIN (V)
4
5
6
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Switch Leakage vs Temperature
VIN = 5.5V RUN = 0V
90 80 70 60 50 40 30 20 10 100 125 0 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125 MAIN SWITCH SYNCHRONOUS SWITCH
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LTC3410 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1 Except for the Resistive Divider Resistor Values) Switch Leakage vs Input Voltage
600 550 500
LEAKAGE CURRENT (pA)
450 400 350 300 250 200 150 100 50 0 0 1 4 3 2 INPUT VOLTAGE (V) 5 6
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MAIN SWITCH
SYNCHRONOUS SWITCH
Start-Up from Shutdown
RUN 2V/DIV VOUT 1V/DIV
IL 200mA/DIV 200s/DIV
VIN = 3.6V VOUT = 1.8V ILOAD = 0A
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Burst Mode Operation
Start-Up from Shutdown
SW 5V/DIV
RUN 2V/DIV
VOUT 50mV/DIV AC COUPLED
VOUT 1V/DIV
IL 100mA/DIV
IL 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 10mA 2s/DIV 200s/DIV
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VIN = 3.6V VOUT = 1.8V ILOAD = 300mA
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Load Step
Load Step
VOUT 100mV/DIV AC COUPLED
VOUT 100mV/DIV AC COUPLED
IL 200mA/DIV
IL 200mA/DIV
ILOAD 200mA/DIV 10s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0mA TO 300mA
ILOAD 200mA/DIV 10s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 20mA TO 300mA
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LTC3410
PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1A supply current. Do not leave RUN floating. GND (Pins 2, 5): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2F or greater ceramic capacitor. VFB (Pin 6): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output.
FU CTIO AL DIAGRA
SLOPE COMP OSC OSC
FREQ SHIFT VFB 6 0.8V
EA
VIN RUN 1 0.8V REF
SHUTDOWN
6
-
IRCMP
+
W
-
+
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0.65V
4 VIN
- +
0.4V
- +
EN SLEEP
-
BURST Q Q SWITCHING LOGIC AND BLANKING CIRCUIT
ICOMP
+
5
S R
RS LATCH
ANTISHOOTTHRU
3 SW
5 2 GND
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LTC3410
OPERATIO
Main Control Loop The LTC3410 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier's output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. Burst Mode Operation The LTC3410 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 70mA regardless of the output load. Each burst event can last from
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(Refer to Functional Diagram)
a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 26A. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier's output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 310kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator's frequency will progressively increase to 2.25MHz when VFB rises above 0V. Dropout Operation As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor.
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LTC3410
OPERATIO
Another important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3410 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section).
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(Refer to Functional Diagram)
Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3410 uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles.
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LTC3410
APPLICATIO S I FOR ATIO
VIN 2.7V TO 5.5V 4.7H CIN 4.7F CER VIN RUN VFB GND 232k 464k
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SW
LTC3410
10pF
Figure 1. High Efficiency Step-Down Converter
The basic LTC3410 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Selection For most applications, the value of the inductor will fall in the range of 2.2H to 4.7H. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is IL = 120mA (40% of 300mA). IL = V 1 VOUT 1- OUT ( f)(L) VIN (1)
The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase.
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Inductor Core Selection
VOUT 1.2V COUT 4.7F CER
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Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3410 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3410 applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER Taiyo Yuden CB2016T2R2M CB2012T2R2M LBC2016T3R3M ELT5KT4R7M CDRH2D18/LD NR30102R2M NR30104R7M FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D MAX DC VALUE CURRENT DCR HEIGHT 2.2H 2.2H 3.3H 4.7H 4.7H 2.2H 4.7H 4.7H 3.3H 2.2H 510mA 530mA 410mA 950mA 450mA 0.13 1.6mm 0.33 1.25mm 0.27 1.6mm 0.2 1.2mm 0.2 2mm
Panasonic Sumida Murata Taiyo Yuden FDK
630mA 0.086 2mm 1100mA 0.1 1mm 750mA 0.19 1mm 1100mA 0.11 1mm 1200mA 0.1 1mm 1300mA 0.08 1mm
LQH32CN4R7M23 4.7H
CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN required IRMS IOMAX
[VOUT (VIN - VOUT )]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher
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LTC3410
APPLICATIO S I FOR ATIO
temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by:
1 VOUT IL ESR + 8fC OUT
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3410's control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires
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can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. The recommended capacitance value to use is 4.7F for both input and output capacitor. For applications with VOUT greater than 2.5V, the recommended value for output capacitance should be increased. See Table 2.
Table 2. Capacitance Selection
OUTPUT VOLTAGE RANGE 0.8V VOUT 2.5V VOUT > 2.5V OUTPUT CAPACITANCE 4.7F 10H or 2x 4.7F INPUT CAPACITANCE 4.7F 4.7F
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Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: R2 VOUT = 0.8V 1+ ( 2) R1 The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 2. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...)
0.8V VOUT 5.5V R2 VFB LTC3410 GND
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R1
Figure 2. Setting the LTC3410 Output Voltage
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LTC3410
APPLICATIO S I FOR ATIO
where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3410 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 3. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages.
1 VIN = 3.6V 0.1
POWER LOSS (W)
0.01
0.001
0.0001
0.00001 0.1
Figure 3. Power Loss vs Load Current
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2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is "chopped" between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3410 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3410 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C,
VOUT = 3.3V VOUT = 1.8V VOUT = 1.2V 10 100 1 LOAD CURRENT (mA) 1000
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LTC3410
APPLICATIO S I FOR ATIO
both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3410 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(JA) where PD is the power dissipated by the regulator and JAis the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: T J = TA + TR where TA is the ambient temperature. As an example, consider the LTC3410 in dropout at an input voltage of 2.7V, a load current of 300mA and an ambient temperature of 70C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70C is approximately 1.0. Therefore, power dissipated by the part is: PD = ILOAD2 * RDS(ON) = 90mW For the SC70 package, the JA is 250C/ W. Thus, the junction temperature of the regulator is: TJ = 70C + (0.09)(250) = 92.5C which is well below the maximum junction temperature of 125C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)).
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Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ILOAD * ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 * CLOAD). Thus, a 10F capacitor charging to 3.3V would require a 250s rise time, limiting the charging current to about 130mA.
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LTC3410
APPLICATIO S I FOR ATIO
PC Board Layout Checklist
When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3410. These items are also illustrated graphically in Figures 4 and 5. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide.
-
VOUT COUT
+
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 4. LTC3410 Layout Diagram
VIA TO GND
VOUT
PIN 1 L1 LTC3410 SW
COUT GND
Figure 5. LTC3410 Suggested Layout
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2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the (-) plates of CIN and COUT as close as possible. 5. Keep the switching node, SW, away from the sensitive VFB node.
1 RUN LTC3410 2 GND VFB VIN 5 CIN VIN 6 R2 3 L1 SW 4 CFWD R1
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R1 VIA TO VIN R2 CFWD VIN VIA TO VOUT
CIN
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LTC3410
APPLICATIO S I FOR ATIO
Design Example
As a design example, assume the LTC3410 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.3A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 3V. With this information we can calculate L using Equation (1), V 1 L= VOUT 1- OUT ( f)(IL ) VIN (3)
Substituting VOUT = 3V, VIN = 4.2V, IL = 100mA and f = 2.25MHz in Equation (3) gives: L= 3V 3V 1- = 3.8H 2.25MHz(100mA) 4.2V
VIN 2.7V TO 4.2V 4 CIN 4.7F CER 3
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 1 10 ILOAD (mA) VIN = 3.6V VIN = 4.2V 100 1000
3410 F06b
Figure 6b
14
U
A 4.7H inductor works well for this application. For best efficiency choose a 350mA or greater inductor with less than 0.3 series resistance. CIN will require an RMS current rating of at least 0.125A ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.5. In most cases, a ceramic capacitor will satisfy this requirement. From Table 2, Capacitance Selection, COUT = 10F and CIN = 4.7F. For the feedback resistors, choose R1 = 301k. R2 can then be calculated from equation (2) to be: V R2 = OUT - 1 R1= 827.8k ; use 825k 0.8 Figure 6 shows the complete circuit along with its efficiency curve.
4.7H* 10pF VIN RUN VFB GND 2, 5 6 825k 301k
3410 F06a
W
UU
SW
VOUT 3V COUT 10F CER TAIYO YUDEN JMK212BJ106 TAIYO YUDEN JMK212BJ475 *MURATA LQH32CN4R7M23

LTC3410 1
Figure 6a
VOUT 100mV/DIV AC COUPLED
IL 200mA/DIV
ILOAD 200mA/DIV 20s/DIV VIN = 3.6V VOUT = 3V ILOAD = 100mA TO 300mA
3410 F06c
Figure 6c
3410f
LTC3410
PACKAGE DESCRIPTIO
0.47 MAX
0.65 REF
2.8 BSC 1.8 REF
RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.10 - 0.40
GAUGE PLANE 0.15 BSC 0.26 - 0.46
NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. DETAILS OF THE PIN 1 INDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE INDEX AREA 7. EIAJ PACKAGE REFERENCE IS EIAJ SC-70 8. JEDEC PACKAGE REFERENCE IS MO-203 VARIATION AB
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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SC6 Package 6-Lead Plastic SC70
(Reference LTC DWG # 05-08-1638)
1.80 - 2.20 (NOTE 4) 1.00 REF 1.80 - 2.40 1.15 - 1.35 (NOTE 4) INDEX AREA (NOTE 6) PIN 1 0.65 BSC 0.15 - 0.30 6 PLCS (NOTE 3) 0.80 - 1.00 0.00 - 0.10 REF 1.00 MAX 0.10 - 0.18 (NOTE 3)
SC6 SC70 1205 REV B
3410f
15
LTC3410
TYPICAL APPLICATIO
VIN 2.7V TO 4.2V 4.7H* 10pF
4 CIN 4.7F
VIN RUN
SW
3
LTC3410 1 6
EFFICIENCY (%)
VFB GND 2, 5
402k 464k
TAIYO YUDEN JMK212BJ475 *FDK MIPF2520D
RELATED PARTS
PART NUMBER LT1616 LT1676 LT1776 LTC1877 LTC1878 LTC1879 LTC3403 LTC3404 LTC3405/LTC3405A LTC3406 LTC3409 LTC3410B LTC3411 LTC3412 LTC3440 DESCRIPTION 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz/2.25MHz, Synchronous Step-Down DC/DC Converter 300mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter with Burst Disabled 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter COMMENTS 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1A, ThinSOT Package 90% Efficiency, VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5A, S8 Package 90% Efficiency, VIN = 7.4V to 40V, VOUT = 1.24V, IQ = 3.2mA, ISD = 30A, N8, S8 Packages 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 15A, ISD = <1A, TSSOP-16 Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable, IQ = 20A, ISD = <1A, DFN Package 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 20A, ISD = <1A, ThinSOT Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20A, ISD = <1A, ThinSOT Package 95% Efficiency, VIN = 1.6V to 5.5V, VOUT = 0.613V, IQ = 65A, DD8 Package 96% Efficiency, VIN = 2.5V to 3.5V, VOUT(MIN) = 0.8V, IQ = 200A, ISD = <1A, SC70 Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60A, ISD = <1A, MS Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60A, ISD = <1A, TSSOP-16E Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25A, ISD = <1A, MS Package
3410f LT 1205 * PRINTED IN USA
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
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Using Low Profile Components, <1mm Height Efficiency
100
VOUT 1.5V COUT 4.7F
3410 TA02
Load Step
VOUT 100mV/DIV AC COUPLED
90 80 70 60 50 40 30 20 10 0 0.1 1 10 ILOAD (mA) VIN = 2.7V VIN = 3.6V VIN = 4.2V 100 1000
3410 TA03
IL 200mA/DIV
ILOAD 200mA/DIV 20s/DIV VIN = 3.6V VOUT = 1.5V ILOAD = 100mA TO 300mA
3410 TA04
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2005

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